Parasitic capacitance cancellation in capacitive measurement

ABSTRACT

An integrated circuit compensates for parasitic capacitance in a capacitive measuring apparatus wherein a capacitance measurement is done by repeatedly transferring charge from a capacitor to be measured to a reference capacitor.

BACKGROUND OF THE INVENTION

This invention relates to the compensation of parasitic or unwanted capacitance in capacitance measurement applications.

Methods of using capacitance measurement to detect the proximity and/or vicinity of an object are known in the art. Inherent parasitic capacitances associated with sense plates, switches, connections and wiring degrade performance in these applications. The same holds true for other capacitance measurement applications. A specific capacitance measurement technique of importance is the “charge transfer method”.

This invention aims to provide a technique to compensate for these parasitic capacitances and thus improve performance.

The invention also relates to an embodiment of the charge transfer method that overcomes the effect of non-linearity in a charging capacitor and enables scaling of a capacitor so that the method can be implemented practically and economically on standard CMOS integrated circuits.

SUMMARY OF THE INVENTION

According to the invention the capacitance measurement is done by repeatedly transferring charge from a capacitor to be measured (C_(M)) to a reference capacitor (C_(R)).

The reference capacitor (C_(R)) starts in a known initial state (e.g. 0V). The measured capacitor C_(M) is charged to a known state (e.g. Vdd). Charge is transferred from C_(M) to C_(R). One such cycle is defined as a “Charge Transfer Cycle”. As more Charge Transfer Cycles are performed the charge, and thus the voltage, on C_(R) increases. The Charge Transfer Cycles continue until C_(R) reaches a specific voltage level (trip level). The time to charge the reference capacitor from the initial state to the trip level is defined as the “measurement period”. The number of transfers in the measurement period is an indication of the size of C_(M).

In each Charge Transfer Cycle an additional unwanted amount of charge is transferred from C_(M) to C_(R) because of parasitic capacitances (C_(P)) in the circuit. This invention relates to the removal of this additional unwanted charge originating from the parasitic capacitances (C_(P)), from C_(R), during each Charge Transfer Cycle so that only the charge from C_(M) remains on C_(R) after a Charge Transfer Cycle. This compensates for the effect of the parasitic capacitances (C_(P)) and only the wanted, measured capacitance on C_(M) is measured.

The sensitivity of the process can be increased by reducing the effective value of C_(M). This means that not only can the parasitic capacitances be removed, but that C_(M) can be adjusted to have a specific capacitance. This must be done without a physical change to the sensor structure (sense plate/antenna). One way to implement this is described hereinafter.

The above capacitance measurement method is still applicable if the reference capacitor (C_(R)) and measured capacitor (C_(M)) are interchanged.

Another charge transfer measurement technique is to perform a set number of Charge Transfer Cycles and measure the final voltage on the reference capacitor C_(R). This method is described in detail in U.S. Pat. No. 7,148,704 by Phillip and in other literature. The same method of compensation for parasitic capacitances (C_(P)) as described above can also be applied for other charge transfer and general capacitive measurement techniques.

In an embodiment, the effect of parasitic capacitance (C_(P)) is cancelled by removing the same amount of charge that the C_(P) added to C_(R) in each Charge Transfer Cycle. Unfortunately for many capacitance measurement methods, the amount of charge the C_(P) and C_(M) capacitors add to C_(R) in each Charge Transfer Cycle is not the same from one Charge Transfer Cycle to the next. Thus the amount of charge that has to be removed from C_(R) to compensate for the effect of the C_(P) capacitor has to change as well. This removal of charge is accomplished by pre-charging a compensation capacitor (C_(C)) and then connecting this C_(C) to the C_(R) capacitor. This connection is such that the net change in voltage of the C_(C) capacitor from the pre-charge value to the final value after connection to C_(R) (ΔV_(Cc)), is the same value but of opposite sign as the change in voltage of C_(P) (ΔV_(CP)) from its pre-charged value (e.g. Vdd) to its final value (e.g. voltage on the C_(R) capacitor (V_(CR))). Thus if C_(P) and C_(C) are of equal value and because the changes in voltage on C_(P) and C_(C) are the same value but of opposite sign (ΔV_(CP)=−ΔV_(Cc)) for each Charge Transfer Cycle, the charge C_(P) adds to C_(R), and the charge C_(C) removes from C_(R), are the same. Thus the effect of the parasitic capacitance (C_(P)) is cancelled.

In another embodiment of the invention, parasitic capacitance cancellation is handled by means of a current mirror structure. A capacitor C_(PC) that defines the parasitic capacitance to be cancelled is charged or discharged after being charged to the same value as C_(M) (e.g. Vdd). The current flowing is then mirrored and extracted from C_(R). This C_(PC) is not necessarily the real parasitic capacitance in the circuit, it is merely a user or designer defined parameter. This has two significant advantages.

Firstly, the current mirror structure is well suited for scaling the current. Thus the reference current (I_(R)) flowing from C_(PC) can be mirrored (I_(R):I_(S)). The notation 1:1 means the exact same current will flow in the secondary circuit; 1:0.5 (or 2:1) gives 50% of the current in the secondary; and 1:2 results in the current in the secondary circuit being doubled. This enables the designer to scale the capacitors, such as C_(PC), in order to optimise for performance, cost, signal to noise ratio or other design parameters.

Secondly, since the charge on C_(R) continuously changes as C_(R) is being charged to the trip level, it is not that simple to remove the correct charge in a constant manner. However, in this embodiment the C_(PC) is charged to a fixed level or discharged from a fixed level to create the reference current. Because of the current mirror operation and characteristics the secondary circuit current (I_(S)) flowing from C_(R) is independent of the voltage level of C_(R). This means the charge added to C_(R) due to the parasitic capacitance can be removed in a way that is constant in every charge/discharge cycle i.e. essentially in the same way that it is added. This provides for a linear operation in removing charge from the C_(R). This, coupled with the implementation described below for handling the charge transfers from C_(M) in the same manner, allows a more linear system is achieved.

In a further embodiment of a charge transfer measurement implementation of capacitive sensing or other sensing (e.g. inductive) the same current mirror structure is used in the discharge cycle of C_(M). So, instead of discharging C_(M) into C_(R) as is common in the art, with the same non-linearity problem due to the rising voltage level in C_(R), C_(M) is discharged through the current mirror to a fixed voltage (e.g. ground). This creates a reference current (I_(R)) and can, again through a current mirror element or similar structure, be used to form a related current (I_(S)) in a secondary circuit. This current can be made to flow into C_(R) transferring the related charge. Due to the characteristics of the current mirror element the current is independent of the voltage level on C_(R) and as such results in the linear charging of C_(R) because the amount of charge transferred is related to the charge flowing from C_(M) and not to the voltage level of C_(R).

In a further embodiment the current mirror or other mirror element can have a ratio between the I_(R) and I_(S) currents. For example if the ratio is a step down ratio of a 1000 then it is possible to reduce the value of C_(R) a 1000 times and still get the same number of transfers. This means a C_(R) that may not be practical to embed in a normal CMOS IC (e.g. 20 nF) can be reduced to a value that can very reasonably be implemented on-chip (e.g. 20 pF). This results in significant savings in cost through reducing discrete components (cost and manufacturing) as well as reducing pin count, or coupling of noise.

Thus, in accordance with the invention, the scaling of charge is used in a charge transfer measurement circuit to facilitate integration of capacitive elements into a standard CMOS integrated circuit.

According to a different aspect of the invention there is provided a parasitic capacitance cancellation circuit in which scaling of charge is effected through a current mirror structure to scale capacitive elements that define or determine the charge which is to be removed from C_(R) during each charge transfer cycle.

The current mirror structure may result in a removal of charge from a reference capacitor in a way that is independent of the voltage level of the reference capacitor.

The invention also extends to a parasitic capacitance cancellation method in which unwanted additional charge which is transferred by a parasitic capacitor to a reference capacitor is removed by a separate compensation capacitor. The term “parasitic capacitance cancellation” is used in a general sense and extends to the cancellation of capacitance inherent to the antenna, sense plate or sensor electrode structure.

Preferably the unwanted additional charge from the parasitic capacitor is removed during each Charge Transfer Cycle. In a variation of the method the unwanted additional charge from the parasitic capacitor produced in one Charge Transfer Cycle is removed during a following Charge Transfer Cycle.

The removal of some of the intrinsic capacitance of the sensor plate is a very powerful technique to enhance or amplify sensitivity. Normally, increasing the size of a sense plate has two opposing effects in that although coupling with an object (say a user hand) is improved, the inherent capacitance of the sense plate is increased. Since a change in capacitance (delta) is measured, the increased inherent capacitance has a negative impact.

In a further embodiment a capacitance cancellation circuit is implemented using the current mirror structure and any applicable ratio for extracting current from C_(R). However, instead of using capacitors to be charged in order to create the reference charge for the cancellation, a reference current is used that flows for a defined period of time. The advantage of this is that it addresses a situation in which extremely small capacitors (femtofarad range) are used in some designs in which the effects of layout (parasitic) capacitance, formed in the lines and as part of the active components, are difficult to plan and simulate.

The capacitance measurement approach can also benefit from an algorithm designed to obtain certain performance objectives. For example, the goal may be sensitivity in which case it would be beneficial to remove as much capacitance as is practical through the capacitance cancellation method or circuit. However, if stability is the main focus then the choice of a larger reference capacitor is better and as such forces a lower capacitance cancellation value. The algorithm needs to take into account the interplay between at least the following group of parameters or subsets thereof: the counts per measurement, the reference capacitor size, the amount of capacitance cancellation, current mirror ratios, the noise in the environment, the transfer frequency, etc.

In practice it has also been found that the use of current mirrors and capacitance cancellation introduces noise into the circuit during some CMOS manufacturing processes. This noise may be described as 1/f noise, Popcorn Noise or Random Telegraph Signal Noise (much is written about this in the literature), and is difficult to remove when present in a process. In general this noise is linked to traps formed in the silicon. For ease of reference this noise is referred to as RTS noise to differentiate it from other random, environmental and system noise.

Because the noise specifically “looks” like or can manifest itself in signals as a genuine signal related for example to a proximity event, it does interfere with operation of the measurement circuit. As such, it is desirable to be able to detect the presence of RTS noise and then to remove the effect of the RTS noise.

To detect the RTS noise it is possible to look for sudden variations or jumps in measurements, but this may coincide with, or may be similar in form to, normal operational events from sense plate signals. However, if the same parts of the measurement circuit (e.g. current mirror and capacitance cancellation circuit) are used to maintain a measurement on the inside of the IC then external events cannot cause a change in the measurement. This internal measurement can then be used to indicate the presence or absence of such noise. In a very simple embodiment this indication of presence of RTS noise may be used to inhibit proximity detections/indications at the time, but still allow touch indications because the RTS noise is not big enough to cause such false measurements.

In a further improvement the change in measurement caused by the RTS noise is quantified and then removed from the measurement signals to yield a “clean” signal on which detection decisions are based. This may be done through analysis of the change in value of the measurement on the internal system, with no external sense plate and only internal components (e.g. fixed capacitors inside the IC for C_(M) and C_(R)), or by monitoring the external signal and measuring changes that occur at the time of RTS noise detection or when the noise falls away. In both cases a quantum of the noise influence on the signal can be determined and the effect thereof removed. In one example such quantum may be added or removed from the long term average to effectively negate its effect in the measurement system.

BRIEF DESCRIPTION OF THE DRAWINGS

The invention is further described by way of examples with reference to the accompanying drawings in which:

FIG. 1 shows a charge transfer circuit with a parasitic capacitance cancellation circuit;

FIG. 2 shows charge (UP) and transfer (PASS) signals;

FIG. 3 depicts a charge transfer circuit and capacitance cancellation circuit during a charge phase;

FIG. 4 depicts the charge transfer and parasitic capacitance cancellation circuit during a transfer phase;

FIG. 5 is a graphical representation of the definition of a mirror structure;

FIG. 6 shows a two stage current mirror structure to transfer charge from C_(M) to C_(R);

FIG. 7 shows a charge transfer circuit which uses a mirror method of parasitic capacitance cancellation;

FIGS. 8A and 8B respectively show two forms of a cascade mirror structure with a current based capacitance cancellation circuit;

FIG. 9 shows a diagram of an integrated circuit using current mirror ratio technology and capacitive cancellation;

FIGS. 10A and 10B are respective flow charts of possible algorithms to implement an automatic adjustment of parameters (CR value or current mirror ratios of amount of capacitive cancellation) for performance based on two specific metrics;

FIG. 11 shows a diagram for a circuit to detect the presence of RTS noise;

FIG. 12 shows an implementation for a capacitive cancellation circuit with the cancellation happening before the current mirror structure; and

FIG. 13 shows a charge scaling capacitor on an input side to reduce the size of ratio required in a current mirror structure when large input capacitors are measured.

DESCRIPTION OF PREFERRED EMBODIMENTS

FIG. 1 shows a charge transfer circuit as well as a parasitic capacitance cancellation circuit.

Before a “measurement period” a reference capacitor (C_(R)) is initialized to a known voltage Va (e.g. 0V) by closing a switch S1.

The Charge Transfer Cycle consists of at least 4 phases, viz the Charge Phase (UP), the Transfer Phase (PASS) and two Deadtime Phases (FIG. 2) which ensure that the Charge Phase and the Transfer Phase do not overlap. During the Charge Phase switches S3, S5 and S7 are closed and switches S2, S4 and S6 are open. A measured capacitor (C_(M)) as well as the parasitic capacitance (C_(P)) are charged to a reference voltage (Vdd).

During the Transfer Phase the switches S₂, S₄ and S₆ are closed and the switches S₃, S₅ and S₇ are open. Capacitors C_(M) and C_(P) are connected to the reference capacitor (C_(R)) and charge is transferred from C_(M) and C_(P) to C_(R). The voltage on C_(R) at the end of the Transfer Phase is V_(CR).

Thus, during each Charge Transfer Cycle, the parasitic capacitance (C_(P)) adds C_(p) (Vdd−V_(CR)) charge to C_(R). This is the amount of charge that the parasitic capacitance cancellation circuit must remove during each Charge Transfer Cycle.

During the Transfer Phase (FIG. 4), a compensation capacitor C_(C) is pre-charged to (Vdd−V_(CR)) and thus C_(C) has a charge of C_(c) (Vdd−V_(CR)). This charge is supplied by the buffer and the supply voltage (Vdd). No charge is added to or removed from C_(R).

During the Charge Phase (FIG. 3) the charge on C_(C) is decreased to 0. The charge needed to change the charge on C_(C) from C_(c) (Vdd−V_(CR)) to 0, is removed from C_(R). The VCc+ terminal of the C_(C) capacitor is connected to a buffer output and the VCc− terminal is connected to C_(R). This connection configuration causes charge to be removed from C_(R) as the C_(C) capacitor discharges from (Vdd−V_(CR)) volt to 0 volt.

Thus if C_(p)=C_(c), the additional charge that the parasitic capacitance (C_(P)) adds to C_(R) during each Charge Transfer Cycle, is removed by the compensation capacitor (C_(C)) in the next Charge Transfer Cycle and the net gain of charge on C_(R) is only because of C_(M). The effect of the parasitic capacitance C_(P) is thus cancelled.

FIG. 5 shows graphically the mirror structure required. The circuit is connected so that a reference current (I_(R)) flows between nodes 1 and 3. The current mirror ratio structure (k defines the ratio between I_(S) and I_(R)) then results in a derived current (I_(S)=kI_(R)) flowing between nodes 2 and 4. k is a factor determined by the designer. If k=1 then the currents are the same; if k<1 then I_(S) is smaller than I_(R), and if k>1, I_(S) is larger than I_(R).

In FIG. 6 the mirror element is shown in a general circuit for charge transfer measurement. C_(M) (the capacitor the value of which must be measured) is charged through S₁ (Charge Phase) and then discharged to ground through S₂ (Transfer Phase). The resulting current (I_(R)) flows through the current mirror element between nodes 1 and 3. This results, in accordance with the method of operation of the mirror element, in a current kI_(R) flowing between nodes 2 and 4. This same current is connected to a second mirror ratio structure and this results in a current n(kI_(R)) flowing between nodes 6 and 8. In this example the initial reference current I_(R) is multiplied first by a factor k and then by a factor n. In a specific situation k and n are each smaller than 1. This current forces charge into C_(R), charging it with a charge that is related through the factor nk with the charge flowing from C_(M) to ground. This technique results in a linear charging of C_(R) i.e. the voltage level on C_(R) rises in equal steps for each Charge Transfer Cycle and does not fall away as results from charging C_(M) directly into C_(R). After each Charge Transfer Cycle the voltage level on C_(R) can be checked to see if a trip level has been reached.

The values of n and k can be chosen to meet certain objectives, for example to limit the value of C_(R) due to size or cost considerations e.g. if k=0.1 and n=0.01 then the charge transfer is reduced by a factor of 1000.

The current mirror can be a single current mirror or use can be made of two or more current mirrors. This does not affect the implementation of the invention. The two stage implementation is merely an example that works well in practice.

FIG. 7 shows the parasitic capacitance cancellation structure. The value of parasitic capacitance to be cancelled can be selectively varied through the various switches available. If the three capacitors shown are used more charge will be removed from C_(R) every cycle.

Through switches S6, S7 and S8 the capacitors are charged. The capacitors are discharged through the switches S₃, S₄ and S₅ to ground, creating a reference current. The mirrored and scaled currents then flow through the secondary nodes of the current mirror structure resulting in charge being removed from C_(R).

During very low voltage levels on C_(R), the structure does not operate well and the charge removed will not reflect the desired parasitic capacitance to be removed. However, in this application it is believed that the negative effect is negligible and is far outweighed by the positives.

The switching selection of the capacitors can be done under software control to automatically calibrate products for optimum operation. For example, a product can be designed and the PCC (parasitic capacitance cancellation) can be used to tune automatically for, say, 4000 transfers, when no touch is present. In this way manufacturing variations can be compensated for.

This means a sensor can be “tuned” to have a certain capacitance and hence a standard level of performance can be achieved over production. It is thus possible to use various current mirror ratios, different size reference capacitors, various capacitance cancellation values and an algorithm to adjust these to obtain specific transfer counts for a fixed trip level with various objectives such as sensitivity (proximity distance), stability, noise immunity, reaction time and number of charge transfer cycles, to reach a specific voltage level (trip level) and sample frequency. These features can be achieved on a single integrated circuit coupled with a sense plate without the need for external capacitors.

The adjustment of a trip level can also be used in an equivalent way to adjusting the C_(R) value. Moving the trip level higher is equivalent to enlarging the C_(R) and vice versa.

FIGS. 8A and 8B show how the charge to be removed during capacitance cancellation can be determined by choosing between various reference current sources (FIG. 8B) rather than capacitors (FIG. 8A). A capacitor charged to a specific voltage contains a defined charge. This charge, divided or multiplied via current mirror ratios, is used to define the charge that is removed in the capacitance cancellation technique. The same effect (FIG. 8B) can be achieved using current as a reference for the charge, instead of capacitance. A defined current flowing for a specific period of time also defines a charge. The charge can be taken out of C_(R) using a current during the complete cycle or during a portion of each cycle. All that is required is that the period (i.e. main oscillator) and amplitude of current remain stable. As mentioned before this may be attractive in terms of implementation on silicon. The charge may also be determined by another technique applicable to the specific situation without affecting the other teachings and advantages of this invention.

Currents are in general more stable and noise immune than voltages. On silicon (CMOS) it is also possible to generate a range of current references, using mirror structures and other techniques, that are well matched and less affected by layout parasitic effect than, for example, capacitors. The use of currents to remove charge from the C_(R) can also have advantages for the capacitive cancellation implementation in the sense that switching every Charge Transfer Cycle is not needed for the cancellation circuit. The cancellation current can flow continuously and as long as the charge transfer frequency is stable a fixed ratio between charge added from the C_(M) to C_(R) and the charge removed from C_(R) will be maintained. If C_(M) changes, the ratio will also change to reflect the changed capacitance measured.

FIG. 9 shows a circuit diagram based on an integrated circuit from Azoteq (Pty) Ltd based on a charge transfer measurement method using current mirror ratios and capacitive cancellation techniques. It is apparent that only a few external components are required and that all C_(R)'s have been implemented on-chip. The implementation of current ratio structures makes the on-chip implementation of components practical and at the same time makes possible the selection and implementation on-chip of at least one such component, multiple components or combinations of such components.

The integrated circuit (U2) (IQS127 from Azoteq (Pty) Ltd) comprises all the building blocks for the capacitive measurement circuit including the current mirror for scaling the charge transferred from the sense plate (which is connected to pad SNS_PLT) to U2 and the capacitive cancellation circuit that contains several capacitors to select from to predetermine the capacitance that is removed. An external resistor R₁ is used to increase protection against electrostatic discharge (ESD) from the sense plate to U2. Capacitors C₁ and C₂ are for voltage regulation and help to assure a good, stable and noise free supply voltage to the IC U2. The device (U2) provides two outputs namely an indication of proximity detection on POUT, and a touch (i.e. much stronger capacitance variation detected) on TOUT.

FIGS. 10A and 10B are two flow charts of respective algorithms for automatic adjustment of parameters to achieve certain performance objectives. The algorithm in FIG. 10A uses the largest acceptable size of C_(R) as a metric to aim for and requires less capacitive cancellation to achieve a certain charge transfer count per cycle. The algorithm in FIG. 10B aims to have the largest acceptable capacitive cancellation amount and this results in a smaller C_(R) value. The FIG. 10B algorithm also results in more sensitive settings for capacitive measurements. The current mirror ratios can also be used to interplay with the C_(R) values or the capacitive cancellation.

Explanation of ATI terms

ATI Antenna Tuning Implementation. C_(R) Reference Capacitor (Four size selections) CC Bits Capacitance Cancellation size selection (0 to 256) Current Sample The number of charge transfers for the current sense channel ATI Target The preselected number of charge transfers that the ATI algorithm aims for CRI_DIV The C_(R) current mirror divider ratio (0 = 32/1 = 128) ATI_BUSY Flag that indicates the ATI is in progress for the current sense channel CRI_DIV Select flag Flag to indicate the C_(R) current mirror divider ratio must be selected CR Select flag Flag to indicate the C_(R) size must be selected ATI_INIT flag Flag to indicate the initial difference (with all the PCC bits set to zero) between the current sample and the ATI target must be stored ATI_AT_MIN flag Flag to indicate the current sample cannot be adjusted lower ATI_AT_MAX flag Flag to indicate the current sample cannot be adjusted higher Long Term Average A filtered value of the current sample Reseed Flag Flag to indicate the long term average must be loaded with the current sample value

Explanation of the ATI Algorithm

The aim of the ATI algorithm is to adjust the relevant parameters (C_(R) size, C_(R) current mirror ratio and the PCC bits) to get the current sample as close as possible to the ATI target count value. This will ensure that the circuit adjusts itself to obtain repeatable performance despite manufacturing and other tolerances.

The ATI algorithm can be implemented in a number of ways. Two possible algorithms are presented. The first algorithm in FIG. 10A (Stability Enhancement) will result in a big C_(R) being selected with a smaller capacitive cancellation (CC) value. This produces a more stable system that is less sensitive and also less noise sensitive. The second algorithm in FIG. 10B (Sensitivity Enhancement) will result in the selection of a big CC value. This produces a more sensitive system that can be used to maximize proximity detection distance.

Algorithm 1 (Stability Enhancement)—FIG. 10A

During initialization for ATI [102], the ATI_BUSY flag is set to indicate to the system that ATI is in progress. The CC bits are set to zero, the current mirror divider ratio is set to a higher value and the C_(R) size is set to a maximum value. The CRI_DIV Select flag is also set to force the system to do a determination if the higher value is the optimal selection.

The system then completes a charge transfer cycle [104]. If it is determined that the CRI_DIV Select flag is set [106] a test is done to check whether the current sample is bigger than the ATI target [138]. If it is bigger the current mirror divider ratio is changed to the lower value [136], the CRI_DIV Select flag is cleared and the CR Select flag is set [134] to force the selection of the appropriate C_(R) size.

After the next charge transfer cycle is completed the CR Select flag is set [108]. The current sample is checked against the ATI target [142]. If the current sample is smaller than the ATI target the CR Select flag is cleared and the ATI_INIT flag is set [140] to start the process of determining the appropriate CC value to get the current sample the closest to the ATI target.

If the current sample is bigger than the ATI target the C_(R) size is reduced [144] to the next smaller value until the current sample is smaller than the ATI target. If the minimum value of C_(R) size is reached [146] the ATI_AT_MIN flag is set [148] to indicate the current sample cannot be adjusted any lower than its current value.

After the next charge transfer cycle is completed the ATI_INIT flag is set [110]. The current CC value (zero) is stored together with the difference between the current sample and the target [134]. The ATI_INIT flag is also cleared.

The algorithm will then keep increasing the CC value [120] and storing the smallest difference value and the CC value that yielded the smallest difference value [118] until either the current sample is at double the target value [112] or the maximum value for the CC is reached [122]. On either of these conditions the CC value that yielded the smallest difference in relation to the target is loaded and a reseed is forced [130].

Algorithm 2 (Sensitivity Enhancement)—FIG. 10B

During initialization for ATI [202], the ATI_BUSY flag is set to indicate to the system that ATI is in progress. The CC bits are set to a third of the maximum value. This will result in the algorithm selecting a smaller C_(R) value with a higher CC value resulting in higher sensitivity. The current mirror divider ratio is set to the lower value and the C_(R) size is set to the maximum value. The CRI_DIV Select flag is also set to force the system to do a determination if the lower value is the optimal selection.

The system then completes a charge transfer cycle [204]. If it is determined that the CRI_DIV Select flag is set [206] a test is done to check whether the current sample is smaller than the ATI target [238]. If it is smaller, the current mirror divider ratio is changed to the higher value [236], the CRI_DIV Select flag is cleared and the CR Select flag is set [234] to force the selection of the appropriate C_(R) size.

After the next charge transfer cycle is completed the CR Select flag will be set [208]. The current sample is checked against the ATI target [242]. If the current sample is smaller than the ATI target the CR Select flag is cleared and the ATI_INIT flag is set [240] to start the process of determining the appropriate CC value to get the current sample the closest to the ATI target. The CC value is also set to zero.

If the current sample is bigger than the ATI target the C_(R) size is reduced [244] to the next smaller value until the current sample is smaller than the ATI target. If the minimum value of C_(R) size is reached the ATI_AT_MIN flag is set [248] to indicate the current sample cannot be adjusted any lower than its current value.

After the next charge transfer cycle is completed the ATI_INIT flag is set [210]. The current CC value (zero) is stored together with the difference between the current sample and the target [234]. The ATI_INIT flag is also cleared.

The algorithm will then keep increasing the CC value [220] and storing the smallest difference value and the CC value that yielded the smallest difference value [218] until either the current sample is at double the target value [212] or the maximum value for the CC is reached [222]. On either of these conditions the CC value that yielded the smallest difference in relation to the target is loaded and a reseed is forced [230]

FIG. 11 illustrates an example of a circuit-noise detection structure which is specifically aimed at noise generated on-chip. An example of the type of noise is Random Telegraph Signal noise (RTS noise) which results in substantial steps in the measurements and which is it not typically Gaussian by nature. The normal implementation incorporates the sense plate, C_(RX) (a reference capacitor for external measurement) and C_(CX) (a reference capacitor for capacitance cancellation of the external sense plate), connected through switches S₁, S₃ and S₅ respectively to a measurement circuit (IC) 320. S₁ is the “PASS” switch in a charge transfer implementation. The “UP” switch is not shown. C_(MI) (internal measurement capacitor) is used to emulate the operation of a sense plate. This is done wholly within the integrated circuit to avoid environmental influences. C_(CI) defines the amount of charge to be removed for the internal measurement. It is important to incorporate as many elements of the circuit as possible for the internal measurement, within the IC.

It is possible but not essential for the internal and external measurements to work concurrently. For example, when one is in the “UP” phase, the other can do the “PASS” phase and vice versa. An additional trip circuit is required for the internal measurements. Detection of a step or change in measurement on the internal C_(RI) indicates a change in the transfer function of the capacitance measurement circuit 320. This is then used for the filtering of the measurement data.

In one embodiment the detection of RTS noise in accordance with the preceding description triggers an analysis of the normal measurement data and an automatic learning algorithm is then implemented to model the noise manifestation from these measurements. It is then possible to remove the effects of this noise automatically from the measurement signal when the noise occurs or when it disappears.

In another embodiment the size (amplitude) of the internal noise is used to derive an effect (through scaling etc) of the noise on the normal measurement and the effect of the noise can be removed.

Various levels of complexity can be involved and this will depend on the requirements in the application and also on the processing resources available to the designers. In a simple form the indication or triggering of a proximity event detection is inhibited for a period when noise is detected.

In analysis it has been found that noise is introduced into the current mirror structures and that when the capacitance cancellation is then performed this noise is amplified. FIG. 12 shows an implementation for capacitive cancellation to reduce or remove the effect of noise amplification when the charge removal is done after the current mirror.

A switch S₁ is a PASS switch that transfers the charge from the Sense Plate (C_(M)) to the current mirror (M1) that mirrors the charge which is transferred as per the ratio (1:X) into the C_(R) where the charge is accumulated to be measured in some way. For example, a fixed trip level may be set and the number of transfers may be counted, or a fixed number of transfers may be done and then the voltage level may be measured with an ND converter.

Essentially the charge from the sense plate is used to change the capacitors C₁ to C_(X), (those connected) before the rest of the charge flows into the current mirror. When S₁ is later opened, a switch S₂ is closed to dump the charge that was accumulated in the capacitance cancellation capacitors. These capacitors must then be charged each time a charge transfer occurs.

In the measurement of small capacitance values the parasitic capacitance inherent in the capacitance cancellation structure may have a negative effect. In this case the structure may be pre-charged (but no cancellation capacitor is switched in) before the charge transfer cycle, to eliminate unwanted parasitic capacitance.

The capacitors C₁ to C_(X) are not effectively used because the input to the current mirror only allows the current mirror to be charged to a threshold value at its input (˜0.7V), whereas the sense plate is charged to a much higher voltage. Hence if these capacitors are pre-charged to a negative voltage it will help to improve size efficiency.

Experience has shown that noise is introduced through the current mirror structures. It has not been determined if higher ratios exacerbate this issue, but in another embodiment (shown in FIG. 13) a simple capacitive charge divider structure is implemented to achieve a scaling effect of the charge transferred from the sense plate (C_(M)) to the measurement circuit. This is important to keep on-chip components, such as capacitors and currents, within practical limits. In FIG. 13 the switch S₂ is closed to charge the sense plate which is effectively a capacitor C_(M). A switch S₂ is then opened and S₁ is closed. This will “pass” the charge from the sense plate to the charge transfer measurement circuit. If S₃ is closed and S₄ is open the charge will be fully transferred and the capacitive cancellation circuit 322 will perform its function on this charge in accordance with its design.

When S₁ is closed, S₃ is open and S₄ is closed the charge from C_(M) is divided between C_(M) and C_(DIV). In the next operation S₁ is opened and S₃ closed. The circuit then operates as before but the charge will have been divided according to the ratio of C_(M) and C_(DIV). When S₁ is opened the process to charge C_(M) through S₂ can start again. It is also preferable that C_(DIV) is chosen so that when the charge division is done, the voltage on C_(DIV) is still higher than the input to the current mirror structure (typically a diode voltage drop). This can also be ensured by not discharging C_(DIV) between each charge transfer cycle. This will change the ratio of charge division but can easily be calculated and accounted for.

The use of the C_(DIV) approach reduces the sensitivity at very high values of C_(M) but provides a large input range. 

1. An integrated circuit for measuring an inductance of an inductive structure, wherein said integrated circuit comprises: charge transfer circuitry for transferring charge during each of a plurality of measurement cycles from said inductive structure to a reference capacitor, and wherein the charge accumulated in the reference capacitor is reduced during every cycle in the inductance measurement by a predetermined amount of charge, and wherein the charge being transferred for measurement is scaled using at least one current mirror structure.
 2. The integrated circuit of claim 1, wherein said integrated circuit further comprises at least one current mirror structure for scaling the amount of charge reduction during every cycle.
 3. The integrated circuit of claim 1, wherein the reference capacitor is located on-chip in the integrated circuit.
 4. The integrated circuit of claim 1 further including a number of selectable on-chip capacitors and wherein the selection of said on-chip capacitors determines the amount of charge reduction during every cycle.
 5. The integrated circuit of claim 1, wherein the current mirror structure or structures includes a selection of scaling ratios that can be selected in accordance with an algorithm to optimize the inductance measurement for sensitivity.
 6. The integrated circuit of claim 1, wherein the current mirror structure or structures includes a selection of scaling ratios that can be selected in accordance with an algorithm to optimize the inductance measurement for speed.
 7. The integrated circuit of claim 1, wherein the current mirror structure or structures includes a selection of scaling ratios that can be selected in accordance with an algorithm to optimize the inductance measurement for stability.
 8. The integrated circuit of claim 2, wherein the current mirror structure or structures includes a selection of scaling ratios that can be selected in accordance with an algorithm to optimize the inductance measurement for sensitivity.
 9. The integrated circuit of claim 2, wherein the current mirror structure or structures includes a selection of scaling ratios that can be selected in accordance with an algorithm to optimize the inductance measurement for speed.
 10. The integrated circuit of claim 2, wherein the current mirror structure or structures includes a selection of scaling ratios that can be selected in accordance with an algorithm to optimize the inductance measurement for stability.
 11. The integrated circuit of claim 2, wherein the reduction of charge is independent of the voltage across said reference capacitor.
 12. The integrated circuit of claim 1, wherein a reference current which flows for a defined period of time is used to reduce said charge by said predetermined amount.
 13. A method to measure an inductance of an inductive structure, wherein said method comprises the steps of transferring charge during each of a plurality of cycles of the measurement from said inductive structure to a reference capacitor, of reducing the amount of charge accumulated in said reference capacitor by a predetermined amount and of scaling the charge being transferred by using at least one current mirror structure.
 14. The method of claim 13, further including a step wherein at least one current mirror structure is used to scale the amount of charge reduction during every cycle of said measurement.
 15. The method of claim 14, further including a step wherein scaling ratios for said current mirror or mirrors are selected according to an algorithm which optimizes said inductance measurement for sensitivity.
 16. The method of claim 14, further including a step wherein scaling ratios for said current mirror or mirrors are selected according to an algorithm which optimizes said inductance measurement for speed.
 17. The method of claim 14, further including a step wherein scaling ratios for said current mirror or mirrors are selected according to an algorithm which optimizes said inductance measurement for stability. 